Method and system for the monolithic integration of circuits for monitoring and control of rf signals

ABSTRACT

A method of operating a BPSK modulator includes receiving an RF signal at the BPSK modulator and splitting the RF signal into a first portion and a second portion that is inverted with respect to the first portion. The method also includes receiving the first portion at a first arm of the BPSK modulator, receiving the second portion at a second arm of the BPSK modulator, applying a first tone to the first arm of the BPSK modulator, and applying a second tone to the second arm of the BPSK modulator. The method further includes measuring a power associated with an output of the BPSK modulator and adjusting a phase applied to at least one of the first arm of the BPSK modulator or the second arm of the BPSK modulator in response to the measured power.

CROSS-REFERENCES TO RELATED APPLICATIONS

This application claims priority to U.S. Provisional Patent ApplicationNo. 61/680,103, filed on Aug. 6, 2012, entitled “Method and System forMonolithic Integration of Circuits for Monitoring and Control of RFSignals,” the disclosure of which is hereby incorporated by reference inits entirety for all purposes.

SUMMARY OF THE INVENTION

Embodiments of the present invention relate to RF circuits. Moreparticularly, embodiments of the present invention relate to anapparatus and method of integrating analog circuits (e.g., monolithicintegration) used to monitor spectral information associated with RFsignals, with particular applications in photonic integrated circuits.

In some embodiments, monolithically integrated circuits are used tomonitor and control RF signals. As described herein, someimplementations utilize analog circuits that are monolithicallyintegrated and are used to monitor the spectrum of RF signals, providingnovel control loops for optical phase shift keying modulationimplementations. Some embodiments use analog circuits to filter out aportion of the spectrum (possibly of a digital signal) that ismonitored, and then an analog circuit, such as a peak detector or RMSdetector to monitor the power in real time.

According to an embodiment of the present invention, a method ofoperating a BPSK modulator is provided. The method includes receiving anRF signal at the BPSK modulator and splitting the RF signal into a firstportion and a second portion that is inverted with respect to the firstportion. The method also includes receiving the first portion at a firstarm of the BPSK modulator, receiving the second portion at a second armof the BPSK modulator, applying a first tone to the first arm of theBPSK modulator, and applying a second tone to the second arm of the BPSKmodulator. The method further includes measuring a power associated withan output of the BPSK modulator and adjusting a phase applied to atleast one of the first arm of the BPSK modulator or the second arm ofthe BPSK modulator in response to the measured power.

According to another embodiment of the present invention, a method ofoperating a QPSK modulator is provided. The method includes receiving anoptical signal at a first BPSK modulator and generating a firstmodulated signal at an output of the first BPSK modulator. The methodalso includes receiving the optical signal at a second BPSK modulatorand generating a second modulated signal at an output of the second BPSKmodulator. The method further includes combining the first modulatedsignal and the second modulated signal at an output of the QPSKmodulator, measuring a power associated with the output of the QPSKmodulator, and adjusting a phase applied to the output of at least oneof the first BPSK modulator or the second BPSK modulator in response tothe measured power.

According to a particular embodiment of the present invention, a nestedMach-Zehnder modulator system is provided. The nested Mach-Zehndermodulator system includes an optical input port and an optical couplercoupled to the optical input port and having a first arm and a secondarm. The nested Mach-Zehnder modulator system also includes a firstinner Mach-Zehnder modulator (MZM) coupled to the first arm. The firstinner MZM comprises a phase control section and an output. The nestedMach-Zehnder modulator system further includes a first optical detectorcoupled to the output of the first MZM, a feedback loop connecting thefirst optical detector to the phase control section, a second inner MZMcoupled to the second arm and having a second phase control section andan output, and a second optical coupler receiving the output of thefirst MZM and the output of the second MZM.

According to a specific embodiment of the present invention, anapparatus is provided. The apparatus includes a receiver coupled to acommunications channel operable to carry an RF signal and a feedbackloop coupled to a transmitter. The apparatus also includes an analogcircuit coupled to the receiver and operable to monitor the RF signal.The analog circuit can include a spectral monitoring unit. The receivercan include a transceiver, which can be integrated in silicon photonics.In some embodiments, the analog circuit includes a plurality of spectralfilters characterized by a differing spectral bands and a plurality ofRMS detectors, each of the RMS detectors being coupled to one of theplurality of spectral filters.

According to another specific embodiment of the present invention, amethod is provided. The method includes receiving an RF signal at aninput of a monolithically integrated analog circuit and measuring the RFsignal using the monolithically integrated analog circuit. The methodalso includes providing a feedback signal based on the measured RFsignal. The monolithically integrated analog circuit can include an RMSdetector. In some embodiments, measuring the RF signal comprisesmeasuring a power associated with the RF signal in one or morepredetermined spectral bands. As an example, the one or morepredetermined spectral bands can be three spectral bands.

Numerous benefits are achieved by way of the present invention overconventional techniques. For example, embodiments of the presentinvention provide methods and systems that avoid heavy digital samplingand processing, which are associated with high power and cost. These andother embodiments of the invention along with many of its advantages andfeatures are described in more detail in conjunction with the text belowand attached figures.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A is a block diagram illustrating a communication system accordingto an embodiment of the present invention.

FIG. 1B is a simplified block diagram illustrating elements of areceiver according to an embodiment of the present invention.

FIG. 2 is simplified block diagram of spectral monitoring unit accordingto an embodiment of the present invention.

FIG. 3 is a plot illustrating filter frequency response curves accordingto an embodiment of the present invention.

FIG. 4 is a plot illustrating the frequency content of an input signalaccording to an embodiment of the present invention.

FIG. 5 is a plot illustrating simulated channel distortion according toan embodiment of the present invention.

FIG. 6 is a plot illustrating RMS detector response for various filtersaccording to an embodiment of the present invention.

FIG. 7 is a simplified schematic diagram illustrating a nested modulatorsystem according to an embodiment of the present invention.

FIG. 8A is a plot illustrating spectral content measured at a detectoraccording to an embodiment of the present invention.

FIG. 8B is a plot illustrating power as a function of modulator biasaccording to an embodiment of the present invention.

FIG. 9 is a simplified block diagram illustrating a peak detectoraccording to an embodiment of the present invention.

FIG. 10 is a plot illustrating the output of a transimpedance amplifieraccording to an embodiment of the present invention.

FIG. 11 is a plot illustrating the peak detector output and the filteroutput according to an embodiment of the present invention.

FIG. 12 is plot illustrating the spectrum of the transimpedanceamplifier according to an embodiment of the present invention.

FIG. 13 is a plot illustrating the spectrum of the filter outputaccording to an embodiment of the present invention.

FIG. 14 is simplified schematic diagram illustrating a nested modulatorsystem according to another embodiment of the present invention.

FIG. 15 is a plot illustrating optical power as a function of time forvarious biases according to an embodiment of the present invention.

FIG. 16 is a plot illustrating integrated signal as a function of biasaccording to an embodiment of the present invention.

FIG. 17 is a simplified circuit diagram for a power detector accordingto an embodiment of the present invention.

FIG. 18 is a plot illustrating transient response at the output of thepower detector according to an embodiment of the present invention.

FIG. 19 is a plot illustrating steady state response at the output ofthe power detector according to an embodiment of the present invention.

FIG. 20 is a plot illustrating voltage output for changes in input noiseaccording to an embodiment of the present invention.

FIG. 21 is a simplified flowchart illustrating a method of operating aBPSK modulator according to an embodiment of the present invention.

FIG. 22 is a simplified flowchart illustrating a method of operating aQPSK modulator according to an embodiment of the present invention.

DETAILED DESCRIPTION OF SPECIFIC EMBODIMENTS

According to an embodiment of the present invention, control loops forquadrature phase shift keying (QPSK) modulators are provided. Incontrast with conventional techniques that sample the signal, perform afast Fourier transform (FFT), and extract a spectral signal to controlthe RF signal, embodiments of the present invention utilize analogcircuits that can be monolithically integrated with photonic circuits,including silicon photonics, to monitor the RF signal and provide thedesired feedback signal. Thus, embodiments of the present inventionreduce the computational complexity, power requirements, cost, and thelike of integrated optoelectronics.

FIG. 1A is a block diagram illustrating a communication system accordingto an embodiment of the present invention. The communications systemprovides methods and systems for signal monitoring and spectrumequalization. Transmitter 110 and receiver 120 are utilized inconjunction with communications channel 115. In the embodimentillustrated in FIG. 1A, the spectral monitoring unit 122 included aspart of the receiver 120 provides feedback on channel distortion to atransmitter 110, thereby allowing the transmitter to compensate for thechannel distortion and correct the spectrum. Feedback from the spectralmonitoring unit 122 is provided using feedback path 130 in theillustrated embodiment.

In a first embodiment of the present invention, an RF signal ismonitored to detect channel distortion and provide feedback to areceiver, in order to equalize the signal spectrum.

FIG. 1B is a simplified block diagram illustrating elements of receiver120 according to an embodiment of the present invention. The receiver120 (e.g., an integrated optical receiver) receives a signal over achannel and couples the signal into a waveguide 150. After thePolarization Diversity Unit (e.g. splitter and rotator) 155 there can betwo waveguides (e.g. for TE polarized light and TM light rotated to TEpolarization). Optical filter 157 is illustrated, which can be a WDMfilter. Light is detected by photodiode 165 and passed to spectralmonitoring unit 160. It should be noted that there can be a plurality ofwaveguides and photodiodes. Also, the methods and systems describedherein can be used outside of optics (i.e., the channel does not have tobe an optical channel). It should be appreciated that the elementsillustrated in FIG. 1B are only exemplary and are not required by thepresent invention and can be utilized, replaced, or removed depending onthe particular application.

FIG. 2 is simplified block diagram of spectral monitoring unit accordingto an embodiment of the present invention. The spectral monitoring unit200 includes a set of three spectral filters: low pass filter (LPF) 220,which passes frequencies less than 4 GHz; band pass filter (BPF) 222,which passes frequencies between 4 GHz and 8 GHz; and high pass filter(HPF) 224, which passes frequencies above 8 GHz. The spectral monitoringunit 200 also includes a set (e.g., three) RMS detectors 230, 232, and234, which receive the outputs of the corresponding filters 220, 222,and 224, respectively. Utilizing the spectral monitoring unit 200, thedistortion as a function of frequency can be measured and feedback canbe provided to the transmitter to compensate for the distortion in thechannel.

Referring to FIG. 2, in order to simulate the channel distortion, the RFsignal is passed through a low pass filter (LPF) 210, which is discussedin relation to FIG. 5. After the signal is received from the channel atreceiver 120, the distorted signal is then provided as an input to thespectral monitoring unit 200, which can include a set (e.g., three)filters that pass predetermined bands. As illustrated, three filters220, 222, and 224 can be utilized: low pass, bandpass, and high pass,but the present invention is not limited to this configuration. Othernumbers of filters with predetermined spectral characteristics can beutilized, for example less than three filters or more than threefilters. Moreover, although frequency ranges in the gigahertz areillustrated, other frequencies can be utilized depending on theparticular application. The outputs from the spectral filters, providingsignals at each of multiple bands, are provided as inputs to a set ofRMS detectors coupled to the filters, providing measurements of thepower in each spectral band. In FIG. 2, the RMS detectors areillustrated as RMS detectors 230, 232, and 234. The outputs of the RMSdetectors provide an indication of the distortion present in thespectrum, and thus can be used to provide feedback for compensation ofthe channel distortion.

FIG. 3 is a plot illustrating filter frequency response curves accordingto an embodiment of the present invention. As can be seen in FIG. 3, theLPF filters signals below 4 GHz, the BPF filters signals between 4 and 8GHz, and the HPF filters signals above 8 GHz. These particular rangesare not required by the present invention and can be modified asappropriate to the particular implementation. One of ordinary skill inthe art would recognize many variations, modifications, andalternatives.

FIG. 4 is plot illustrating the frequency content of an input signalaccording to an embodiment of the present invention. For this inputsignal, which is associated with a 12 Gpbs PRBS signal, a dip can beseen in the spectrum at 12 GHz.

In an example, the V_(RF) _(—) _(in) shown in FIG. 2 is the voltage of aRF signal, and in particular it may be a pseudo random bit series (PRBS)at a desired bit rate (e.g., 12 Gbps), useful for demonstrating theoperation of the methods and systems described herein.

FIG. 5 is a plot illustrating simulated channel distortion according toan embodiment of the present invention. As illustrated in FIG. 5,channel distortion can be modeled and/or simulated by a Low Pass Filter(LPF) with an attenuation of 10 dB between 1 GHz and 12 GHz.

FIG. 6 is a plot illustrating RMS detector response for various filtersaccording to an embodiment of the present invention. Referring to FIG.6, the output of the set of RMS detectors is illustrated as “RMSDetector Response,” showing the difference between the output, forexample, of the low pass filter with no channel distortion (LPF) and theoutput of the low pass filter with the channel distortion applied (LPFch). Curves for the other spectral filters show similar decreases in RMSdetector output when channel distortion is introduced—BPF and BPF ch;HPF and HPF ch. As illustrated in FIG. 6, the RMS detector output islower for each of the filtered signals when channel distortion isincluded.

Thus, the use of the analog detector (e.g., an RMS detector) to measurethe output power in the various spectral bands enables a measurement ofthe impact of the introduction of channel distortion (represented by adecrease in output response in this example), which can then be used toprovide a feedback signal to compensate for the channel distortion.Because the analog circuits can be monolithically integrated, they canbe used in place of digital circuits that could be used to provide thefeedback control signals. As an example, if no channel distortion ispresent, the received signal may be flat as a function of frequency,resulting in equal voltages at each of RMS detectors 230, 232, and 234.When channel distortion is present, attenuation at some frequencies canresult in a decrease in the voltage measured at one or more of the RMSdetectors (e.g., attenuation of low frequencies in comparison to highfrequencies). The relative voltage decrease, which is seen to be highestat the HPF, where the signal was attenuated the most, can provide dataused in the feedback loop to the transmitter to compensate for thechannel distortion and return the RMS voltages to a common level. One ofordinary skill in the art would recognize many variations,modifications, and alternatives.

In summary, embodiments provide spectral monitoring of a digital signalsuch that channel distortion can be detected and feedback provided forsignal equalization. In an embodiment, a PRBS at 12 Gbps is generatedand represented by ±0.15 V. Referring to FIG. 2, the spectrum can befiltered, using analog filters, into three spectral bands. The signals,after filtering, are each passed to one of a set of RMS detectors, whichare used to measure the power present in each spectral band. Assuminguniform signals as a function of frequency, if the signal is distortedby the channel, then the output power at each RMS detector will change.Thus, the distortion resulting from the channel will be detected and canbe used to provide feedback for the equalization of the signal attransmission.

FIG. 7 is a simplified schematic diagram illustrating a nested modulatorsystem according to an embodiment of the present invention. This nestedmodulator system allows for control of null bias in an opticalMach-Zehnder Modulator (MZM) as described below.

As described more fully below, the embodiment discussed in relation toFIG. 7 includes a design for a control loop to maintain the bias pointof an optical MZM. As a specific example, this control technique can beused to maintain the null bias in a Binary Phase Shift Keying (BPSK)modulation format. In addition to BPSK, this control technique can beused for the inner modulators in a nested MZM configuration for QPSK,DQPSK, or DP-QPSK. One of ordinary skill in the art would recognize manyvariations, modifications, and alternatives.

Referring to FIG. 7, the nested MZM configuration includes two innerMZMs, first MZM 710 and second MZM 712 connected together within anouter MZM 700. MZM 700 includes an input port 705 and an output port atnode 730. For QPSK, each MZM uses a specific bias point for operation:

The bias point of the inner MZMs 710 and 712 is maintained such that theoutput of the inner MZMs when the arms are not driven otherwise, is aminimum (null bias). The two high-speed RF inputs (e.g. RF-I andRF-I_(bar)) drive the two arms (e.g. with diodes) to obtain two outputsthat are π phase shifted from each other.

The bias point of the outer MZM 700 is maintained such that the outputof the inner MZMs are 90° out of phase (i.e., π/2) with one another(quadrature bias), as discussed below.

The control loop for null bias relies on the application of two pilottones that produce intermodulation tones in the output signal of theMZM. A peak detector analog circuit is then used to monitor theintensity of the odd intermodulation frequencies, and provide thefeedback to maximize this quantity and maintain the null bias.

Referring to FIG. 7, the first inner MZM 710 receives I and I_(bar)signals and the second inner MZM 712 receives the Q and Q_(bar) signalsfor the I and Q constellation of the QPSK. Phase control is provided toeach inner MZM and is illustrated by Phase 1+Pilot 1/Phase 2+Pilot 2provide to the upper and lower arms of the inner MZMs as shown in FIG.7. The power at node 720 is measured using monolithically integratedmonitor photodetector (MPD 1), the power at node 722 is measured usingMPD2, and the power at node 730 is measured using MPD3. Phase 3 is usedto modify the phase of the lower arm of the outer MZM 700. A variety ofphase adjustment devices can be utilized including heaters integratedwith waveguide elements, electro-optic phase adjustment elementsincluding diode-based phase adjustment sections, and the like. It isunderstood that in each case, only the phase difference between the twoarms is of importance, and the actual phase shifters can be applied toone or both of the arms of each MZ (inner or outer).

FIG. 8A is a plot illustrating spectral content measured at a detectoraccording to an embodiment of the present invention. For thisembodiment, Pilot Tone 1 (f₁) was set at 3 kHz and Pilot Tone 2 (f₂) wasset at 4 kHz. FIG. 8A shows the spectrum measured at MPD1, with peaks inthe power spectrum at 2f₁ (6 kHz), 2f₂ (8 kHz), f₁+f₂ (7 kHz), andf₁−f₂(1 kHz).

FIG. 8B is a plot illustrating power as a function of modulator biasaccording to an embodiment of the present invention. FIG. 8B displaysthe inter-modulation power at f₁−f₂, which is related to the nonlineartransfer function of the MZM. By measuring the power in theinter-modulation spectrum, a feedback signal is provided that can beused to control-loop the phase shift, preferably at π in this example.The phase can be adjusted until the inter-modulation spectrum ismaximized, providing the desired null (π) bias point.

In the embodiment illustrated in FIG. 7, power measurements are madeusing the MPD 1 and MPD 2 (which detect intensity) to determine that a πphase shift is present between the two arms of each MZM 710 and 712. Inthe illustrated example, MPD1 is used to control MZM 710 by measuringthe power at the inter-modulation frequency of 1 kHz. Although 3 kHz and4 kHz are utilized for the pilot tones in this example, the presentinvention is not limited to these frequencies and other suitablefrequencies can be utilized as appropriate to the particularapplication. The pilot tones can be applied to separate arms or can beapplied to the same arm. Although not illustrated in FIG. 7 for purposesof clarity, the power measured at MPD 2 can be used to control Phase 2in the lower arm of MZM 712. Thus, the description that is provided inrelation to MPD 1 and MZM 710 is also applicable, as appropriate, to MPD2 and MZM 712. One of ordinary skill in the art would recognize manyvariations, modifications, and alternatives.

Referring once again to FIG. 8B, the desired π bias is correlated to thepower of the inter-modulation tone (f₁−f₂) and by maximizing this tone,the desired π bias can be maintained.

Thus, the control loop will maximize this tone in this implementation toachieve the desired null bias on MZMs 710 and 712. It should be notedthat these same techniques can be used for control of individual BPSKMZMs. Thus, although a QPSK configuration is illustrated in FIG. 7, thisis not required by the present invention and this null bias techniquecan be implemented for a single BPSK modulator as well. One of ordinaryskill in the art would recognize many variations, modifications, andalternatives.

FIG. 9 is a simplified block diagram illustrating a peak detectoraccording to an embodiment of the present invention. The exemplaryanalog circuit (i.e., peak detector circuit) illustrated in FIG. 9 canbe used to follow the power of the signal at the output of the MZMs,including MZM 710, MZM 720, or MZM 700 in real time in order to maximizethe inter-modulation tone. The monitor photodiode 910 measures the powerof the output of the MZM, which is amplified using transimpedanceamplifier (TIA) 912 with resistor 913 in parallel, filtered using theband pass filter (BPF) 914, passed to the peak detector 916, whichincludes a diode 917, and capacitor 918, which measures the peak power.Thus, by monitoring the intensity using the MPDs illustrated in FIG. 7,which may include the peak detector shown in FIG. 9, the signal appliedto the phase shifter (e.g. Phase 1 or Phase 2 in FIG. 7) can be modifiedto maximize the measured signal and thus the null bias. In anembodiment, phase shifters can be implemented using thermal devices(e.g., heaters) or other suitable systems to adjust the phase in the MZMarms.

FIG. 10 is a plot illustrating the output of a transimpedance amplifieraccording to an embodiment of the present invention. The output of TIA912 oscillates as a function of time as illustrated. FIG. 11 is a plotillustrating the peak detector output and the filter output according toan embodiment of the present invention. As shown in FIG. 11, the outputof BPF 914 shows a reduced oscillation amplitude, and number ofoscillations (as only the low frequency of interest is kept), incomparison to the TIA output. The output of peak detector 916 issubstantially constant as it is associated with the voltage on the peakdetector capacitor 918. Thus, using the peak detector illustrated inFIG. 9, the MPDs are able to provide a measure of the RMS power at thevarious nodes as illustrated.

FIG. 12 is plot illustrating the spectrum of the transimpedanceamplifier according to an embodiment of the present invention.

FIG. 13 is a plot illustrating the spectrum of the filter output (BPF914 as illustrated in FIG. 9) according to an embodiment of the presentinvention. As illustrated in FIG. 13, a peak is present at 1 kHz, whichis associated with the difference frequency between the two pilot tones.

FIG. 14 is a simplified schematic diagram illustrating a nestedmodulator system according to another embodiment of the presentinvention. In the nested modulator system illustrated in FIG. 14, whichshares common elements with the nested modulator system shown in FIG. 7,the power measured at node 1405 is utilized to adjust the phase appliedto the modulator arm by Phase 3. Within the QPSK modulation format, twoindependent pseudo-random data streams are applied to the two innerMZMs. The application of the two independent data streams (which aretypically uncorrelated) produces broadband “noise” at the output of theouter MZM when the quadrature bias is not maintained. As illustrated inFIG. 14, an RMS analog circuit 1410 is used in conjunction with monitorphotodiode MPD 3 to detect the power and reduce or minimize the noisegenerated and to maintain the quadrature bias.

The “noise” is correlated to the phase error in relation to the desiredphase shift of n/2. Accordingly, a portion of the broadband noise issampled and reduced or minimized to achieve the desired phase shift. Asillustrated, MPD 3 can be used to control the bias associated with Phase3, providing for quadrature on the outer MZM. Thus, in this embodiment,the integrated AC time signal power or the integrated power spectrum isreduced or minimized to control the quadrature bias.

Mathematically, the control of the quadrature bias through reductions inthe broadband noise can be described as follows:

E = ^(−j ϕ₁(t)) + ^(−j ϕ₂(t))^(−j ϕ_(b)), where${\phi_{1} = {I\; \frac{\pi}{2}}},{\phi_{2} = {Q\; \frac{\pi}{2}}},{{{and}\mspace{14mu} \phi_{b}} = {\frac{\pi}{2\;} + {\Delta \; \phi_{b}}}},$

where φ₁ is the relative phase at node 1420 (after the first inner MZM),φ₂ is the relative phase at node 1422 (after the second inner MZM),φ_(b) is the bias applied by Phase 3 and Δφ_(b) is the error in thephase, and E is the combined electrical field at 1405 after the innerMZMs coherently combine.The power measured at MPD 3 is proportional to:

P_(D)(t) = EE^(*) = 2[1 + cos (ϕ₁ − ϕ₂ − ϕ_(b))]${P_{D}(t)} = {2{\left\{ {1 + {\sin \left\lbrack {{\left( {I - Q} \right)\frac{\pi}{2}} - {\Delta \; \phi_{b}}} \right\rbrack}} \right\}.}}$

The AC component is

${P_{D,{A\; C}}(t)} = {2{{\sin \left\lbrack {{\left( {I - Q} \right)\; \frac{\pi}{2}} - {\Delta \; \phi_{b}}} \right\rbrack}.}}$

Since I and Q can only receive the values ±1 we have

$\left( {I - Q} \right) = \left\{ \begin{matrix}0 \\{- 2} \\{2,}\end{matrix} \right.$

and then for all three cases P_(D,AC)(t)=0 if Δφ_(b)=0. Thus, byadjusting Phase 3, the power measured at MPD 3 can be driven towardszero as Δφ_(b)=0.

FIG. 15 is a plot illustrating optical power as a function of time forvarious biases according to an embodiment of the present invention. Forthe bias equal to 0.95π/2, the optical power associated with the “noise”in the data ranges from about 0.46 to about 0.54 on a normalized powerbasis (i.e., arbitrary units). As the bias is adjusted closer to the π/2objective, i.e., to 0.97π/2, the optical power decreases to a range ofabout 0.48 to about 0.52. For a bias level at the desired π/2, theoptical power amplitude fluctuation associated with the noise isnegligible. Thus, reductions in the noise level towards a constant (DC)output power can be used to determine if the bias is at the desiredvalue.

FIG. 16 is a plot illustrating integrated signal as a function of biasaccording to an embodiment of the present invention. The noise power(i.e., the integrated signal over a 1 GHz bandwidth) 1605 is plotted asa function of bias normalized to π/2. Although some embodiments utilizean RF signal with a wide bandwidth (e.g., 30 GHz), a portion of the wideband noise (e.g., 1 GHz) can be used according to an embodiment of thepresent invention. By measuring the noise level and reducing the noiselevel, it is possible to provide better than 1% control over thequadrature bias, which corresponds to less than 0.5°. Although a “noise”power bandwidth of 1 GHz is utilized in FIGS. 15 and 16, this is notrequired by the present invention and other bandwidths can be utilized.

Curve 1610 in FIG. 16 represents the noise level resulting from othersystem elements, providing a floor for minimization. Thus, embodimentsof the present invention can reduce the noise associated with the phaseerror to less than the noise floor from other system elements.

FIG. 17 is a simplified circuit diagram for a power detector accordingto an embodiment of the present invention. The exemplary RMS circuitillustrated in FIG. 17 can receive the filtered noise signal. The outputindicates the power present in the filtered noise band.

For the power detector illustrated in FIG. 17, if the input signal is ACcoupled, squared, and low pass filtered, it is possible to obtain anoutput voltage that is proportional to the square of the RMS voltage.

For an input signal:

${V\left( {\omega \; t} \right)} = {\frac{a_{0}}{2} + {\left\lbrack {{\sum\limits_{n = 1}^{\infty}{a_{n}{\sin \left( {n\; \omega \; t} \right)}}} + {\sum\limits_{n = 1}^{\infty}{b_{n}{\cos \left( {n\; \omega \; t} \right)}}}} \right\rbrack.}}$

If the input signal is AC coupled and the result squared:

V ²(ωt)=k[Σ _(n=1) ^(∞) a _(n) sin(nωt)+Σ_(n=1) ^(∞) b _(n) cos(nωt)]².

After low pass filtering, the output voltage is:

$V_{out} = {{\frac{k^{\prime}}{2}\left\lbrack {{\sum\limits_{n = 1}^{\infty}a_{n}^{2}} + {\sum\limits_{n = 1}^{\infty}b_{n}^{2}}} \right\rbrack} = {{k^{\prime}{\sum\limits_{n = 1}^{\infty}V_{{{RM}\; S},n}^{2}}} = {k^{\prime}{V_{{RM}\; S}^{2}.}}}}$

FIG. 18 is a plot illustrating transient response at the output of thepower detector according to an embodiment of the present invention. Fora phase error of −1% to +1%, the output voltage is represented by thecurves 1810 and 1812, respectively, which are characterized by a smallamplitude. As the phase error increases to ±5% and ±10%, the outputvoltage increases as illustrated by curve 1820 (±5%) and curves 1830(−10%) and 1832 (+10%). As illustrated in FIG. 18, the transientresponse demonstrates that fast changes can be accomplished.

FIG. 19 is a plot illustrating steady state response at the output ofthe power detector according to an embodiment of the present invention.For an error of about 10%, the steady state voltage is about 0.6 V(curve 1930 for −10% and curve 1932 for +10%). As the error decreases,the voltage drops as shown by curve 1920 at ˜0.13 V for ±5% and curve1910 at ˜0 V for ±1%.

FIG. 20 is a plot illustrating voltage output for changes in input noise(i.e., abrupt change in the phase bias error) according to an embodimentof the present invention. When the quadrature error was increased from5% to 10% at a time of about 17 μs, the voltage changed from the steadystate value of ˜0.15 V to a new steady state value of ˜0.6 V in a periodof ˜10 μs. For systems using thermal control to modify the phase values,time constants of external stimuli are typically on the order ofmilliseconds. For these systems, the response times of microsecondsillustrated in FIG. 20 are fully suitable.

FIG. 21 is a simplified flowchart illustrating a method of operating aBPSK modulator. The method includes receiving an RF signal at the BPSKmodulator (2110) and splitting the RF signal into a first portion and asecond portion that is inverted with respect to the first portion(2112). As illustrated in FIG. 7, the RF I and Q signals are split intoinverted signals and applied to the BPSK modulators. The method alsoincludes receiving the first portion at a first arm of the BPSKmodulator (2114) and receiving the second portion at a second arm of theBPSK modulator (2116).

The method further includes applying a first tone to the first arm ofthe BPSK modulator (2118) and applying a second tone to the second armof the BPSK modulator. Additionally, the method includes measuring apower associated with an output of the BPSK modulator (2122) andadjusting a phase applied to at least one of the first arm of the BPSKmodulator or the second arm of the BPSK modulator in response to themeasured power (2124). Adjusting the phase can include increasing thepower associated with the output of the BPSK modulator, for example,maximizing the power. In some embodiments, measuring the powerassociated with the output of the of the BPSK modulator comprisesspectrally filtering the output, for example, by performing band passfiltering at a difference frequency of the first tone and the secondtone.

It should be appreciated that the specific steps illustrated in FIG. 21provide a particular method of operating a BPSK modulator according toan embodiment of the present invention. Other sequences of steps mayalso be performed according to alternative embodiments. For example,alternative embodiments of the present invention may perform the stepsoutlined above in a different order. Moreover, the individual stepsillustrated in FIG. 21 may include multiple sub-steps that may beperformed in various sequences as appropriate to the individual step.Furthermore, additional steps may be added or removed depending on theparticular applications. One of ordinary skill in the art wouldrecognize many variations, modifications, and alternatives.

FIG. 22 is a simplified flowchart illustrating a method of operating aQPSK modulator. The method includes receiving an optical signal at afirst BPSK modulator (2210) and generating a first modulated signal atan output of the first BPSK modulator (2212). In an embodiment, theoptical signal is a cw signal provided by a laser source. The methodfurther includes receiving the optical signal at a second BPSK modulator(2214) and generating a second modulated signal at an output of thesecond BPSK modulator (2216).

The method also includes combining the first modulated signal and thesecond modulated signal at an output of the QPSK modulator (2218) andmeasuring a power associated with the output of the QPSK modulator(2220). As illustrated in FIG. 14, measuring the power can include theuse of an RMS analog circuit. Also, measuring the power can includespectrally filtering the output of the QPSK modulator into one or morespectral bands before passing the filtered signal to detectors, forexample, by applying a 1 GHz bandwidth bandpass filter to the output.Additionally, the method includes adjusting a phase applied to theoutput of the second BPSK modulator in response to the measured power(2222). It should be understood that it is only the phase differencebetween the two BPSK modulators is of importance. Thus the phase can beadjusted after each one of the BPSK modulators, or on both of them.

As illustrated in FIG. 16, adjusting the phase applied to the output ofthe second BPSK modulator can include increasing the phase toward a π/2bias, resulting in a decrease in power as the integrated signalapproaches the minimum at π/2 bias. Moreover, adjusting the phaseapplied to the output of the second BPSK modulator can includedecreasing the phase toward a π/2 bias, resulting in a decrease inpower. The phase adjustment can be accomplished in several ways,including adjusting a temperature of a heater associated with the secondBPSK modulator.

It should be appreciated that the specific steps illustrated in FIG. 22provide a particular method of operating a QPSK modulator according toan embodiment of the present invention. Other sequences of steps mayalso be performed according to alternative embodiments. For example,alternative embodiments of the present invention may perform the stepsoutlined above in a different order. Moreover, the individual stepsillustrated in FIG. 22 may include multiple sub-steps that may beperformed in various sequences as appropriate to the individual step.Furthermore, additional steps may be added or removed depending on theparticular applications. One of ordinary skill in the art wouldrecognize many variations, modifications, and alternatives.

It is also understood that the examples and embodiments described hereinare for illustrative purposes only and that various modifications orchanges in light thereof will be suggested to persons skilled in the artand are to be included within the spirit and purview of this applicationand scope of the appended claims.

What is claimed is:
 1. A method of operating a BPSK modulator, themethod comprising: receiving an RF signal at the BPSK modulator;splitting the RF signal into a first portion and a second portion thatis inverted with respect to the first portion; receiving the firstportion at a first arm of the BPSK modulator; receiving the secondportion at a second arm of the BPSK modulator; applying a first tone tothe first arm of the BPSK modulator; applying a second tone to thesecond arm of the BPSK modulator; measuring a power associated with anoutput of the BPSK modulator; and adjusting a phase applied to at leastone of the first arm of the BPSK modulator or the second arm of the BPSKmodulator in response to the measured power.
 2. The method of claim 1wherein adjusting the phase comprises increasing the power associatedwith the output of the BPSK modulator.
 3. The method of claim 2 whereinincreasing the power comprises maximizing the power.
 4. The method ofclaim 1 wherein measuring the power associated with the output of theBPSK modulator comprises spectrally filtering the output.
 5. The methodof claim 4 wherein spectrally filtering the output comprises performingband pass filtering at a difference frequency of the first tone and thesecond tone.
 6. The method of claim 1 wherein measuring the powerassociated with the output of the BPSK modulator comprises using a peakdetector.
 7. A method of operating a QPSK modulator, the methodcomprising: receiving an optical signal at a first BPSK modulator;generating a first modulated signal at an output of the first BPSKmodulator; receiving the optical signal at a second BPSK modulator;generating a second modulated signal at an output of the second BPSKmodulator; combining the first modulated signal and the second modulatedsignal at an output of the QPSK modulator; measuring a power associatedwith the output of the QPSK modulator; and adjusting a phase applied tothe output of at least one of the first BPSK modulator or the secondBPSK modulator in response to the measured power.
 8. The method of claim7 wherein measuring the power comprises using an RMS analog circuit. 9.The method of claim 7 wherein measuring the power comprises spectrallyfiltering the output of the QPSK modulator.
 10. The method of claim 9wherein spectrally filtering comprises applying a 1 GHz bandwidthbandpass filter to the output.
 11. The method of claim 7 whereinadjusting the phase difference between the two BPSK modulators comprisesincreasing the phase toward a π/2 bias, resulting in a decrease inpower.
 12. The method of claim 7 wherein adjusting the phase differencebetween the two BPSK modulators comprises decreasing the phase toward aπ/2 bias, resulting in a decrease in power.
 13. The method of claim 7wherein adjusting the phase difference between the two BPSK modulatorscomprises adjusting a temperature of a heater associated with one ormore of the BPSK modulators.
 14. A nested Mach-Zehnder modulator systemcomprising: an optical input port; an optical coupler coupled to theoptical input port and having a first arm and a second arm; a firstinner Mach-Zehnder modulator (MZM) coupled to the first arm, wherein thefirst inner MZM comprises a phase control section and an output; a firstoptical detector coupled to the output of the first MZM; a feedback loopconnecting the first optical detector to the phase control section; asecond inner MZM coupled to the second arm and having a second phasecontrol section and an output; and a second optical coupler receivingthe output of the first MZM and the output of the second MZM.
 15. Thenested MZM system of claim 14 further comprising a third phase controlsection coupled to at least one of the output of the first MZM or theoutput of the second MZM.
 16. The nested MZM system of claim 15 furthercomprising a third optical detector coupled to an output of the secondoptical coupler.
 17. The nested MZM system of claim 16 furthercomprising a second feedback loop connecting the third optical detectorto the third phase control section.
 18. The nested MZM system of claim14 further comprising a second optical detector coupled to the output ofthe second MZM.